The present invention relates to a high voltage stabilizing circuit which generates stabilized high voltage from commercial alternating-current power to provide anode voltage to be applied to an anode electrode of a cathode-ray tube, for example.
Television receivers, projectors, and display units for personal computers which employ cathode-ray tubes (hereinafter referred to as CRTs) as display devices have widely spread.
As is well known in the art, in various display units having such CRTs, a required level of high voltage (anode voltage) needs to be supplied in a stable manner to anode electrodes of the CRTs.
Specifically, although the level of anode voltage is generally set at about 25 KV to 35 KV, the anode voltage is varied according to variations in alternating-current input voltage and a load, for example. Variations in the anode voltage may in turn vary screen sizes in a vertical and a horizontal direction of an image displayed on the CRT. In addition, variations in the anode voltage cause a bright white peak image to be distorted when displayed on the screen. As shown in FIG. 9 for example, when an original rectangular white peak image as indicated by a solid line is displayed, the white peak image is distorted into a trapezoidal shape as indicated by a broken line.
Therefore, in practice, a so-called high voltage stabilizing circuit is provided as an anode voltage supplying circuit to stabilize anode voltage for output.
FIG. 6 is a circuit diagram showing an example of a high voltage stabilizing circuit. The high voltage stabilizing circuit shown in the figure is to be provided for a so-called multiscanning-capable display unit for a personal computer.
First, FIG. 6 shows a rectifying and smoothing circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. The rectifying and smoothing circuit rectifies and smoothes commercial alternating-current power AC to provide a rectified and smoothed voltage Ei whose level is equal to that of the commercial alternating-current power AC. The rectified and smoothed voltage Ei is supplied as direct-current input voltage to a switching power supply circuit 10.
The switching power supply circuit 10 is a DC-to-DC converter configured to provide a direct-current output voltage Eo by switching and stabilizing the inputted rectified and smoothed voltage Ei. In this case, the switching power supply circuit 10 outputs a direct-current output voltage Eo stabilized at 240 V.
The direct-current output voltage Eo is inputted to a step-down converter 20.
The step-down converter 20 is for example formed by connecting a drain and a source of a MOS-FET switching device Q11 in series with a choke coil CH1 between a line of the direct-current output voltage Eo and a positive electrode of a smoothing capacitor COA and inserting a diode DD1 between a point where the source of the switching device Q11 and the choke coil CH1 are connected and a primary-side ground.
The switching device Q11 is externally driven by a driving voltage from a driving circuit 14, which will be described later, to perform switching on the direct-current output voltage Eo. A current that flows according to switching operation by the switching device Q11 is stored in the smoothing capacitor COA via the choke coil CH1 and the diode DD1. Then, the step-down converter 20 outputs a stepped-down direct-current voltage EOA, which is a voltage across the smoothing capacitor COA.
The stepped-down direct-current voltage EOA is supplied to a voltage resonance type converter 30.
The voltage resonance type converter 30 shown in FIG. 6 is provided with a MOS-FET switching device Q12 to perform externally excited single-ended operation.
In the voltage resonance type converter 30, a drain of the switching device Q12 is connected to a positive terminal of the stepped-down direct-current voltage EOA via a choke coil CH2, while a source of the switching device Q12 is connected to the primary-side ground. A gate of the switching device Q12 is supplied with a driving voltage outputted from a driving circuit 16, which will be described later. The switching device Q12 is driven for switching operation by this driving voltage.
The drain of the switching device Q12 is also connected to a starting point of a primary winding N1 of a flyback transformer FBT, which will be described later. In this case, an ending point of the primary winding N1 is grounded to the primary-side ground via a direct current blocking capacitor C11.
Thus, switching output of the switching device Q12 is transmitted to the primary winding N1 of the flyback transformer FBT, whereby an alternating voltage in accordance with the switching frequency is obtained at the primary winding N1.
A parallel resonant capacitor Cr is connected in parallel with the drain and source of the switching device Q12. Capacitance of the parallel resonant capacitor Cr, inductance L12 of the choke coil CH2, and leakage inductance L1 of the primary winding N1 side of the flyback transformer FBT form a primary-side parallel resonant circuit of the voltage resonance type converter. Although detailed description is omitted here, voltage V2 across the resonant capacitor Cr actually forms a sinusoidal pulse waveform during an off period of the switching device Q12 as a result of the action of the parallel resonant circuit, so that voltage resonance type operation is obtained.
Also, a clamp diode DD2 is connected in parallel with the drain and source of the switching device Q12. The clamp diode DD2 forms a path of clamp current that flows during the off period of the switching device Q12.
The switching devices Q11 and Q12 on the primary side are driven for switching operation by the following configuration.
On the basis of a horizontal synchronizing signal frequency fH corresponding to a currently set resolution, a synchronizing circuit 11 generates and outputs a horizontal synchronizing signal having the frequency fH. Since the high voltage stabilizing circuit in this case is to be provided for a multiscanning display unit, the horizontal synchronizing signal frequency fH is varied within a range of 30 KHz to 120 KHz, for example.
In this case, the horizontal synchronizing signal generated by the synchronizing circuit 11 is inputted to an oscillating circuit 12. The oscillating circuit 12 converts the horizontal synchronizing signal into an oscillating frequency signal to be used for driving the switching devices Q11 and Q12, and outputs the converted signal to a PWM control circuit 13 and a driving circuit 16.
The driving circuit 16 generates a driving voltage for driving the switching device Q12 from the inputted oscillating frequency signal, and outputs the driving voltage to the gate of the switching device Q12. Thus, the switching frequency of the switching device Q12 coincides with the currently set horizontal synchronizing signal frequency fH.
The PWM control circuit 13 effects PWM control of the inputted oscillating frequency signal on the basis of a detection output from an error amplifier circuit 15. Specifically, the PWM control circuit 13 variably controls a duty ratio of an on/off period in one cycle of the oscillating frequency signal, and outputs the result to a driving circuit 14. The driving circuit 14 generates driving voltage using a PWMed oscillating frequency signal outputted from the PWM control circuit 13, and outputs the signal to the switching device Q11. Thus, the switching frequency of the switching device Q11 also coincides with the currently set horizontal synchronizing signal frequency fH. This means that both the switching device Q11 and the switching device Q12 are allowed to perform switching operation at a switching frequency in synchronism with the horizontal synchronizing signal frequency fH. It should be noted that a duty ratio of an on/off period in one switching cycle of the switching device Q12 is governed by the waveform (duty ratio in one cycle) of the oscillating frequency signal changed by the PWM control circuit 13. This is related to stabilization of the level of high direct-current voltage EHV, that is, control for constant voltage, which will be described later.
The switching output obtained at the switching device Q12 of the voltage resonance type converter 30 described above is supplied to a high voltage generating circuit 40.
The high voltage generating circuit 40 comprises the flyback transformer FBT and a high voltage rectifier circuit provided on a secondary side thereof. The switching output of the switching device Q12 is transmitted to the primary winding N1 of the flyback transformer FBT.
As shown in FIG. 6, the primary winding N1 is wound on the primary side of the flyback transformer FBT. Also, five step-up windings NHV1, NHV2, NHV3, NHV4, and NHV5 are wound as secondary windings on the secondary side of the flyback transformer FBT. In practice, each of these step-up windings NHV1 to NHV5 is wound around a core in a divided and independent state. The step-up windings NHV1 to NHV5 are wound in such a manner that their polarity is opposite to that of the primary winding N1, whereby flyback operation is obtained.
As shown in FIG. 6, the step-up windings NHV1, NHV2, NHV3, NHV4, and NHV5 are connected in series with high voltage rectifier diodes DHV1, DHV2, DHV3, DHV4, and DHV5, respectively, thereby forming a total of five half-wave rectifier circuits. The five half-wave rectifier circuits are connected in series with each other. Then, a smoothing capacitor COHV is connected in parallel with the five serially connected half-wave rectifier circuits.
Thus, on the secondary side of the flyback transformer FBT, the five half-wave rectifier circuits rectify voltages induced in the step-up windings NHV1 to NHV5 to store the rectified voltages in the smoothing capacitor COHV. Accordingly, the level of direct-current voltage across the smoothing capacitor COHV is five times as high as the voltage induced in each of the step-up windings NHV1 to NHV5. The direct-current voltage across the smoothing capacitor COHV is converted into high direct-current voltage EHV to be used as anode voltage for a CRT, for example.
Operation for constant voltage by the circuit shown in FIG. 6 will next be described.
A series connection circuit of a voltage dividing resistance R1-R2 is connected in parallel with the smoothing capacitor COHV at both terminals thereof provided on the secondary side of the flyback transformer FBT. Hence, a point of connection between the voltage dividing resistances R1 and R2 has a voltage level obtained by dividing high direct-current voltage EHV by their voltage dividing ratio. The point of connection between the voltage dividing resistances R1 and R2 is connected to an input of the error amplifier 15. The error amplifier 15 compares the level of the divided high direct-current voltage EHV with a predetermined reference level to detect an error. Specifically, the error amplifier circuit 15 detects an error of the high direct-current voltage EHV level with respect to the predetermined reference level. Then, the error amplifier circuit 15 outputs a direct current or a direct-current voltage whose level is changed according to the amount of the error, for example.
The detection output of the error amplifier 15 is supplied to the PWM control circuit 13. The PWM control circuit 13 effects PWM control of the inputted oscillating frequency signal on the basis of the detection output from the error amplifier 15, and outputs the result to the driving circuit 14.
Thus, the switching device Q11 driven by the driving circuit 14 is fixed at the switching frequency in synchronism with the horizontal synchronizing signal frequency, and performs switching operation in accordance with the on/off-period duty ratio that is changed by the PWM control according to variations in the level of the high direct-current voltage EHV.
The level of stepped-down direct-current voltage EOA is determined by the level of the direct-current output voltage Eo, the switching frequency of the switching device Q11, and the duty ratio of an on/off period in one switching cycle of the switching device Q11. The stepped-down direct-current voltage EOA is expressed as:
EOA=Eoxc2x7TON1/Ts
where one cycle of switching operation by the switching device Q11 is Ts and an on period in one cycle is TON1.
When the duty ratio of an on/off period in one switching cycle of the switching device Q11 is changed by the PWM control according to an error of the high direct-current voltage EHV level, as described above, the level of the stepped-down direct-current voltage EOA can be variably controlled.
FIGS. 7A to 7H show operating waveforms of the switching devices Q11 and Q12 in the circuit shown in FIG. 6. FIGS. 7A to 7D show operations at a horizontal synchronizing signal frequency fH=30 KHz, while FIGS. 7E to 7H show operations at the same points as in FIGS. 7A to 7D respectively at a horizontal synchronizing signal frequency fH=120 KHz.
FIGS. 7A and 7E show drain-to-source voltage V1 of the switching device Q11, and FIGS. 7B and 7F show switching current I1 that flows through the drain of the switching device Q11.
FIGS. 7C and 7G show voltage V2 across the parallel resonant capacitor Cr (parallel resonance voltage) connected in parallel with the switching device Q12, and FIGS. 7D and 7H show switching current I2 that flows through the drain of the switching device Q11.
FIGS. 7A and 7B show operations under PWM control at a horizontal synchronizing signal frequency fH=30 KHz where one switching cycle Ts of the switching device Q11 is 3.3 xcexcs, and the period TON1 during which the switching device Q11 is turned on is 6.9 xcexcs. In this case, the level of the stepped-down direct-current voltage EOA is about 50 V.
FIGS. 7E and 7F show operations under PWM control at a horizontal synchronizing signal frequency fH=120 KHz where one switching cycle Ts of the switching device Q11 is 8.3 xcexcs, and the period TON1 during which the switching device Q11 is turned on is 0.4 xcexcs. In this case, the level of the stepped-down direct-current voltage EOA is about 220 V.
As indicated by operations of the switching device Q12 shown in FIGS. 7C and 7D and FIGS. 7G and 7H, in the voltage resonance type converter 30 powered for operation by the stepped-down direct-current voltage EOA, a period TOFF2 during which the switching device Q12 is turned off is set to be constant at 3 xcexcs irrespective of change in the horizontal synchronizing signal frequency fH. The period TOFF2 can be set to be constant by selecting the capacitance of the parallel resonant capacitor Cr.
When effecting PWM control of switching operation by the switching device Q11 according to variation in the level of the high direct-current voltage EHV and setting the off period TOFF2 of the switching device Q12 to be constant at 3 xcexcs as described above, it is possible to control the parallel resonance voltage V2 to a constant level irrespective of changes in the horizontal synchronizing signal frequency fH, as indicated in FIG. 7C.
The parallel resonance voltage V2 is transmitted as switching output to the primary winding N1 of the flyback transformer FBT. Therefore, when the parallel resonance voltage V2 is controlled to a constant level, an alternating voltage obtained at the primary winding N1 of the flyback transformer FBT is also maintained at a constant level. Accordingly, the high direct-current voltage EHV generated on the secondary side of the flyback transformer FBT is also controlled to a constant level.
The high voltage stabilizing circuit shown in FIG. 6 thus stabilizes the level of the high direct-current voltage EHV.
As a summary of the operations described above, characteristics of variations of the high direct-current voltage EHV, the direct-current output voltage Eo, and the stepped-down direct-current voltage EOA with respect to the horizontal synchronizing signal frequency fH will be shown in FIG. 8.
According to the figure, the direct-current output voltage Eo is stabilized at 240 V by stabilizing operation of the switching power supply circuit 10.
The stepped-down direct-current voltage EOA is controlled so as to change linearly in a range of about 50 V to 220 V for horizontal synchronizing signal frequencies fH=30 KHz to 120 KHz. In FIG. 8, stepped-down direct-current voltages EOA when a load current IHV supplied by the high direct-current voltage EHV to a load is 1 mA and 0 mA are shown. Although the stepped-down direct-current voltage EOA at a load current IHV=1 mA is somewhat lower than the stepped-down direct-current voltage EOA at a load current IHV=0 mA, substantially the same voltage value is obtained.
Thus, by controlling the level of the stepped-down direct-current voltage EOA as described above and by setting the parallel resonance voltage V2 generated by the switching operation of the switching device Q12 to be constant irrespective of change (switching) of the horizontal synchronizing signal frequency fH as described above, the high direct-current voltage EHV becomes constant at about 27 KV, for example.
The high voltage stabilizing circuit shown in FIG. 6 thus stabilizes the high direct-current voltage EHV to suppress variations in screen sizes in a vertical and a horizontal direction of an image displayed on the CRT.
In the case of the high voltage stabilizing circuit configured as shown in FIG. 6, the trapezoidal distortion of the bright rectangular white image, which is caused by a variation xcex94EHV in the high direct-current voltage EHV due to an increase in high voltage load current IHV occurring at the white peak, is dealt with by increasing the capacitances of the smoothing capacitors CO, COA, and COHV to a required level.
However, the high voltage stabilizing circuit configured as shown in FIG. 6 has the following problems.
First, the high voltage stabilizing circuit configured as shown in FIG. 6 performs power conversion in two stages by means of the step-down converter 20 and the voltage resonance type converter 30 in the process from the input of commercial alternating-current power to the generation of the high direct-current voltage EHV. Thus, overall power conversion efficiency in the high voltage stabilizing circuit as a whole is low; for example, the power conversion efficiency at a high voltage load power PHV=27 W (=high direct-current voltage EHVxc3x97IHV) is actually no more than 72%, and reactive power in this case is about 10.5 W. Then input power becomes correspondingly greater.
Moreover, with such a configuration, the actually assembled circuit also has a complex structure, thus resulting in increase in the number of parts.
Specifically, for example, the circuit requires two ferrite transformers (an FBT and an insulating converter transformer in the switching power supply circuit 10), two ferrite choke coils (CH1 and CH2), and at least three switching devices (Q11, Q12, and a switching device in the switching power supply circuit 10). Therefore, the size of the circuit itself becomes large as a result of large mounting area of its printed board, for example, and also the cost of the circuit becomes correspondingly high. In addition, the size and cost of parts in the circuit are increased because the capacitances of the smoothing capacitors CO, COA, and COHV are increased to suppress the trapezoidal distortion of the bright rectangular white image, as described above.
In view of the problems described above, an object of the present invention is to provide a high voltage stabilizing circuit which improves power conversion efficiency and reduces the size and cost thereof.
To achieve the above object, according to an aspect of the present invention, there is provided a high voltage stabilizing circuit including rectifying and smoothing means for rectifying and smoothing commercial alternating-current power and thereby providing direct-current input voltage; switching means having a switching device for interrupting the direct-current input voltage for output; a parallel resonant circuit on a primary side formed so as to convert operation of the switching means into voltage resonance type operation; a high voltage output transformer having a primary-side winding to which switching output of the switching means is transmitted and a secondary-side winding closely coupled to the primary winding for inducing a required high level of stepped-up alternating voltage; high direct-current voltage generating means supplied with the stepped-up alternating voltage for generating and outputting a required high level of direct-current voltage; and constant voltage control means for controlling the high direct-current voltage to a constant voltage by changing switching frequency of the switching device according to level of the high direct-current voltage.
As described above, the high voltage stabilizing circuit according to the present invention directly transmits switching output of a voltage resonance type converter provided on the primary side to the primary side of the high voltage output transformer, or for example a flyback transformer. A high direct-current voltage is obtained on the secondary side of the high voltage output transformer by using the stepped-up alternating voltage induced by the switching output on the primary side. Also, the high direct-current voltage is stabilized actively by variably controlling the switching frequency of the primary-side voltage resonance type converter according to the level of the high direct-current voltage.
Such a configuration requires only one voltage resonance type converter to serve as a switching converter for power conversion of direct-current input voltage obtained on the primary side from commercial alternating-current power.
The above and other objects, features and advantages of the present invention will become apparent from the following description and the appended claims, taken in conjunction with the accompanying drawings in which like parts or elements denoted by like reference symbols.